Method for the secondary error correction of a multi-port network analyzer

ABSTRACT

A method for the error correction of a vectorial network analyzer, where a primary system calibration is initially implemented using a calibration kit. Following this, a first, secondary error correction is implemented on at least two one-port networks of the vectorial network analyzer. After this first, secondary error correction of the one-port networks of the vectorial network analyzer, a second, secondary error correction is implemented, where either two one-port networks are through-connected in an ideal manner or a measurement is implemented on a reciprocal two-port network. The corrected system-error values from the first, secondary error correction are used even in this further measurement, and overall, a high-precision, calibrated multi-port network analyzer is obtained.

CROSS-REFERENCE TO RELATED APPLICATION

The present application claims priority to German Application No. 10 2009 024 751.3, filed on Jun. 12, 2009, the entire contents of which are herein incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates to a method for the secondary error correction of a vectorial multi-port network analyzer.

2. Discussion of the Background

For the accurate measurement of complex-value scattering parameters with a vectorial network analyzer (VNA), it is initially necessary to implement a system calibration in the measuring planes of the network analyzer. The measuring planes are generally the ends of the test cables. A plurality of different calibration methods are known for this purpose. TOSM, TRL, LRL, LRM, which differ from one another with regard to the necessary calibration standards and the subsequent evaluation of the individual measurements, can be named as examples. Dependent upon the calibration method and the number of test ports, a measurement of different standards (no-load, short-circuit, broadband load, sliding load, (air-) lines, direct through-connection of the test ports) is carried out, from which a determination of the system error parameters is implemented. Using the system error parameters, a numerical error correction is performed in the subsequent object measurement. This is known, for example, from DE 39 12 795 A1.

After the completion of the error correction, the magnitude of the residual error parameters, which can also be described as effective system parameters, is primarily determined, in addition to the methodology of the calibration method and the care taken with the calibration procedure, through the accuracy of the description of the calibration standards, which the manufacturer of a calibration kit supplies for the user. With the introduction of electronic calibration kits, but also with the miniaturisation associated with relatively higher frequencies of mechanical calibration standards, the direct reference to the mechanical properties of the calibration standard has been lost, so that the user is unrestrictedly dependent upon the manufacturer's information. In addition to the system calibration by means of the named methods, a verification method is therefore often also used, or a secondary calibration is implemented.

In order to estimate the measurement errors based on a potentially error-laden calibration, it is proposed in regulation EA-10/12, that the absolute value of the effective directivity and of the effective source-port match be determined by connecting a coaxial precision air line incorrectly terminated or short-circuited in a defined manner at its output to the test port to be measured of the system-calibrated network analyzer. In order to determine the effective directivity and the effective source-port match, the absolute value of the reflection coefficient at the input of this air line is measured within the ( . . . ) test frequency range. With the oscillations of the absolute value of the reflection coefficient observed in this test configuration, the oscillation amplitude (also referred to as the ripple amplitude) provides a measure for the absolute value of the effective directivity or respectively of the effective source-port match. However, this method provides only a relatively imprecise estimate of the effective system parameters, from which a re-correction cannot be derived.

A method for determining the effective directivity and effective source-port match present at a test port of a calibrated network analyzer which is improved compared to the above-described is known from WO 03/076956 A2. This method is also based on the measurement of a precision air line short-circuited at the output-end and provides very accurate and moreover complex values for the effective directivity and effective source-port match, and in fact for every frequency point within the test frequency range. The values of the complex reflection coefficient measured within a fine frequency raster are subjected to a numerical frequency-offset, extrapolation, inverse Fourier transform, and the searched effective system parameters are extracted by low-pass filtering from the reflectometry signal obtained accordingly in the time domain. With the determination of the complex-value, effective directivity and source-port match, a re-correction can then be implemented, and the measurement accuracy can therefore be increased. Furthermore, it is possible to determine the complex-valued effective reflection synchronization and accordingly the third system parameter required for the complete description of one-port measurements.

The disadvantage with the known method is that a secondary error correction can only be used on one-port networks.

SUMMARY OF THE INVENTION

Accordingly, embodiments of the present invention advantageously provide a method for reducing the residual error in a vectorial multi-port network analyzer.

The present invention relates to a method for secondary error correction of a multi-port network analyzer, wherein a correction comprises all system-error parameters between the test ports (in transmission), therefore also the correction of those additionally relevant in the case of a two-port measurement. As already explained, it is known that a determination of the effective source-port match and effective directivity and their re-correction for all one-port networks of a network analyzer can be implemented, through the implementation of an air line measurement, the subsequent numerical determination of the effective directivity and of the effective source-port match and the accordingly possible calculation of their respectively corrected values separately at every port of the VNA. Starting from such a partial secondary error correction, according to the invention, a further measurement, this time of the two-port network, is performed. In this context, either a through-connection (as ideal as possible) is established or a reciprocal network is inserted between the two test ports. Both approaches have in common that a further measurement is implemented with the system parameters of the one-port networks having already been subjected to a partial secondary correction.

In the case of the ideal through-connection, after the determination on the one-port networks of at least one of the system parameters directivity, source-port match and reflection synchronization, a second, secondary error correction is performed, which comprises a measurement of the scattering parameter matrix on at least one frequency point. From this scattering parameter matrix, an un-corrected scattering parameter matrix is determined, from which finally a corrected load-port match and/or a corrected transmission tracking is determined.

In the case of the reciprocal network, a transmission matrix is measured in order to determine the corrected transmission tracking.

Since a secondary one-port error correction has already been performed in each case with high precision before the measurement of the two-port network, a correction of the system error parameters relevant for the transmission measurement (load-port match, transmission tracking) can be performed with the measurement now to be implemented on the two-port network.

In particular, the determination of the scattering matrix or the transmission matrix in each case for all frequency points of a measurement range can be determined, for which corrected system parameters for the one-port networks are available. Moreover, in addition to the already described determination of the system parameters directivity and source-port match, it is also advantageous to determine the reflection synchronization. For this purpose, a further measurement is performed according to the invention on a short-circuited or offset short-circuited air line.

It is particularly advantageous to perform the described second secondary error correction in pairs for different test ports, thereby calibrating the multi-port network analyzer as a whole in a high-precision manner.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is explained in greater detail below with reference to schematic drawings and diagrams. The individual drawings are as follows:

FIG. 1 shows a signal flow diagram for an individual VNA test port;

FIG. 2 shows a signal flow diagram for a VNA test port, which is terminated with an air line short-circuited at the output end;

FIG. 3 shows a measured absolute value |Γ_(m)| of the input reflection coefficient of an air line short-circuited at the end;

FIG. 4 shows a Fourier transform of the test signal Γ_(m) according to FIG. 3 with characteristic maxima;

FIG. 5 shows a 12-term system error model of a vectorial 2-port network analyzer in forward direction;

FIG. 6 shows a 12-term system error model of a vectorial 2-port network analyzer in reverse direction;

FIG. 7 shows an 8-term system error model of a vectorial 2-port network analyzer; and

FIG. 8 shows a simplified method flow-chart of the method according to the invention.

DETAILED DESCRIPTION OF EXEMPLARY EMBODIMENTS OF THE INVENTION

In the mathematical description below, the meaning of the symbols used is as follows:

-   A directivity term -   a₀ incident wave at port 1 -   a′₃ incident wave at port 2 -   α attenuation constant of the reference air line -   B short-circuit term -   b₀ reflected wave at port 1 -   b′₀ transmitted wave at port 1 -   b₃ transmitted wave at port 2 -   b′₃ reflected wave at port 2 -   C source-match term -   c speed of light -   ε₀₀ effective directivity (residual directivity) -   e₀₀ directivity {tilde over (e)}₀₀ error-corrected directivity -   ε₁₁ effective source-port match (residual source match) -   e₁₁ source-port match -   {tilde over (e)}₁₁ error-corrected source-port match -   ε₁₀ε₀₁ effective reflection synchronization (residual reflection     synchronisation) -   e₁₀e₀₁ reflection synchronization -   {tilde over (e)}₁₀{tilde over (e)}₀₁ error-corrected reflection     synchronization -   e₁₀e₃₂ transmission tracking -   {tilde over (e)}₁₀{tilde over (e)}₃₂ error-corrected transmission     tracking -   e₂₂ load-port match in forward direction -   {tilde over (e)}₂₂ error-corrected load-port match in forward     direction -   γ propagation constant on the air line -   Γ_(a) actual reflection coefficient of the device under test (DUT) -   Γ_(m) measured reflection coefficient after the system-error     correction -   {tilde over (Γ)}_(m) frequency-offset, measured reflection     coefficient -   Γ_(a) ^(SC) model for the offset short-circuit -   Γ_(m) ^(SC) measurement of the offset short-circuit -   λ attenuation factor -   L length of the air line including offset short-circuit line portion -   Φ phase factor -   S_(ij) scattering parameters of the air line including offset     short-circuit line portion -   [S_(T)] measured scattering matrix of a through-connection of two     test ports -   [S_(T,raw)] raw value scattering matrix corresponding to the matrix     [S_(T)] -   ω angular frequency

The signal flow diagram for the individual ports of a VNA as presented in FIG. 1 is initially assumed.

Accordingly, the measured reflection coefficient Γhd m and the actual value Γ_(a) for the reflection coefficient of the device under test DUT relate to one another according to

$\begin{matrix} {\Gamma_{m} = {\frac{b_{0}}{a_{0}} = {ɛ_{00} + {\left( {ɛ_{10}ɛ_{01}} \right){\frac{\Gamma_{a}}{1 - {ɛ_{11}\Gamma_{a}}}.}}}}} & (1) \end{matrix}$

As already mentioned, the reflection coefficients and the system parameters are complex-valued, frequency dependent values. On the assumption that the effective system error parameters are small, that is to say, a primary calibration of the VNA has been performed, equation (1) can be approximated by

Γ_(m)≈ε₀₀+(ε₁₀ε₀₁)Γ_(a)+ε₁₁(ε₁₀ε₀₁)Γ_(a) ²  (2).

FIG. 2 shows the signal flow diagram for the measurement to be performed after the primary calibration for the first, secondary error correction on the one-port networks, in which an air line short-circuited at the output end is connected to the test port, wherein the short-circuit is generally performed as an offset short-circuit. The wave propagation on the air line including offset short-circuit can be described by the complex propagation constant

$\begin{matrix} {{\gamma = {\alpha + {j\frac{\omega}{c}}}},} & (3) \end{matrix}$

wherein the attenuation constant is assumed to be small by comparison with the phase constant. Moreover, ideal cross-sectional dimensions for the air line are assumed, so that the following applies for their reflection scattering parameters

S₁₁=S₂₂=0  (4)

and their input reflection coefficient can be approximated by

Γ_(a) ≈−S ₂₁ S ₁₂ =−e ^(−2γL) =−λe ^(−j2ωL/c)  (5).

The factor

λ=e^(−2αL)  (6)

describes the small attenuation losses on the air line or respectively the offset short-circuit.

By inserting (5) into (2), the following is obtained for the measured reflection coefficients:

$\begin{matrix} {{\Gamma_{m} \approx {{\underset{\underset{A}{}}{ɛ_{00}}\underset{\underset{B}{}}{{- \left( {ɛ_{10}ɛ_{01}} \right)}{\lambda }^{{- j}\; 2\omega \; {L/c}}}} + \underset{\underset{C}{}}{{ɛ_{11}\left( {ɛ_{10}ɛ_{01}} \right)}\lambda^{2}^{{- j}\; 4\omega \; {L/c}}}}},} & (7) \end{matrix}$

of which the summands can be assigned to the effective directivity (A), to a short-circuit term (B) and to source-port match term (C).

It is evident that the measured reflection coefficient of a short-circuited air line provides phase displacements −2ωL/c and 4ωL/c rising in a linear manner with the frequency. While the directivity term (A) remains unchanged, the short-circuit term (B) is modulated with double, and the source-port match term (C) with four-times the phase displacement caused by the propagation of the microwave on the air line. These frequency dependencies deviating from one another by the factor two cause the typical oscillations in the absolute value of the reflection coefficient |Γ_(m)| in the case of a VNA calibrated in a non-ideal manner, as shown in FIG. 3.

Provided the length of the air line does not fall below a minimum dimension, the Fourier transform of the test signal Γ_(m) shows the characteristic presented in FIG. 4 in the time domain, in which three mutually separated, characteristic maxima are recognisable, which can be assigned to the terms (A), (B) and (C).

It is assumed that the residual error terms are slowly variable with reference to their frequency dependency by comparison with the phase factor

Φ:=e^(−j2ωL/c)  (8)

and accordingly, an extraction of the characteristic maxima in FIG. 4 is possible by filtering in the time domain. Since a low-pass filtering with fixed filter bandwidth is possible with the method because of the equidistant spacing of the partial reflections in the time domain, a numerical downward mixing through division in each case by integer multiples of the phase factor Φ is implemented before the time domain transformation, in each case if required. After the low-pass filtering, a re-transformation into the frequency domain is performed with subsequent upward mixing through multiplication by the phase factor Φ.

Explicit reference is made to details of the determination of the effective source-port match and of the effective directivity of a one-port network, as described in WO 03/076956 A2. However, the determination of the residual reflection synchronisation (δ₁₀ε₀₁) should be additionally implemented here, and accordingly, the determination of all residual reflection terms is explained below. For this purpose, the test signal is numerically extrapolated at both limits of the test-frequency range before the time-domain transformation by linear prediction, which is performed with the present method in the same manner as in WO 03/076956 A2 and will not be described again here.

The determination of the terms (A), (B) and (C) in equation (7) begins with the largest component (B) with regard to absolute value by implementing a numerical downward mixing into the baseband according to

$\begin{matrix} {{\overset{\sim}{\Gamma}}_{m} = {{\Gamma_{m}/\Phi} \approx {{ɛ_{00}^{j\; 2\omega \; {L/c}}} - {\left( {ɛ_{10}ɛ_{01}} \right)\lambda} + {{ɛ_{11}\left( {ɛ_{10}ɛ_{01}} \right)}\lambda^{2}{^{{- j}\; 2\omega \; {L/c}}.}}}}} & (9) \end{matrix}$

After {tilde over (Γ)}_(m) has been extrapolated according to the above method, a low-pass filtering of (ε₁₀ε₀₁)λ in the time domain is performed, from which the term (B) is obtained after re-transformation into the frequency domain and inverse frequency displacement. Since the attenuation factor λ is not known sufficiently accurately, the residual reflection synchronisation (ε₁₀ε₀₁) cannot be determined directly. However, the term (B) is required for the further calculation. In the next step, the term (A), that is, the residual directivity ε₀₀, is determined by low-pass filtering of

Γ_(m) −B≈A+C=ε ₀₀+ε₁₁(ε₁₀ε₀₁)λ² e ^(−j4ωL/c)  (10).

For the calculation of the term (C), equation (10) is converted according to

$\begin{matrix} {{\frac{\Gamma_{m} - B}{B^{2}} \approx {\frac{A}{B^{2}} + \frac{C}{B^{2}}}} = {{\frac{ɛ_{00}}{\left( {ɛ_{10}ɛ_{01}} \right)^{2}\lambda^{2}}^{j\; 4\omega \; {L/c}}} + \frac{ɛ_{11}}{\left( {ɛ_{10}ɛ_{01}} \right)}}} & (11) \end{matrix}$

and once again, the scheme of extrapolation, time-domain transformation and low-pass filtering is applied. In this context, not only the phase rotation Φ² is eliminated, but also the unknown attenuation factor λ². The expression

$\begin{matrix} {\frac{C}{B^{2}} = \frac{ɛ_{11}}{\left( {ɛ_{10}ɛ_{01}} \right)}} & (12) \end{matrix}$

obtained from this in the frequency domain, ignoring the residual reflection synchronisation ((ε_(1o)ε₀₁)≈1), is equal to the sought residual source-port match.

The hitherto unknown residual reflection synchronisation (ε_(1o)ε₀₁) is determined through an additional measurement of an offset short-circuit. In fact, according to equation (1), between the measurement Γ_(m) ^(SC) and the model Γ_(a) ^(SC) of the offset short-circuit obtained from simulations, the relationship

$\begin{matrix} {\Gamma_{m}^{SC} = {ɛ_{00} + {\left( {ɛ_{10}ɛ_{01}} \right)\frac{\Gamma_{a}^{SC}}{1 - {ɛ_{11}\Gamma_{a}^{SC}}}}}} & (13) \end{matrix}$

must be fulfilled. Accordingly, the following is obtained for the residual reflection synchronisation

$\begin{matrix} {\left( {ɛ_{10}ɛ_{01}} \right) = {\left( {{\frac{C}{B^{2}}\Gamma_{a^{*}}^{SC}} + \frac{\Gamma_{a}^{SC}}{\Gamma_{m}^{SC} - A}} \right)^{- 1}.}} & (14) \end{matrix}$

With the knowledge of the complex reflection synchronisation, the residual source-port match ε₁₁ according to equation (12) can be determined more accurately.

With the complex-valued residual error parameters determined preferably for every test frequency point for the directivity ε₀₀, the source-port match ε₁₁ and the reflection synchronization (ε₁₀ε₀₁), a secondary error correction can be performed at every test port of the VNA. The corrected system error parameters are calculated from the error parameters previously obtained through the system calibration and the now known, complex residual error parameters as follows:

$\begin{matrix} {{{\overset{\sim}{e}}_{00}e_{00}} + {\left( {e_{10}e_{01}} \right)\frac{ɛ_{00}}{1 - {e_{11}ɛ_{00}}}}} & (15) \\ {{\overset{\sim}{e}}_{11} = {ɛ_{11} + {\left( {ɛ_{10}ɛ_{01}} \right)\frac{e_{11}}{1 - {e_{11}ɛ_{00}}}}}} & (16) \\ {{{\overset{\sim}{e}}_{10}{\overset{\sim}{e}}_{01}} = {\left( {ɛ_{10}ɛ_{01}} \right)\frac{e_{10}e_{01}}{\left( {1 - {e_{11}ɛ_{00}}} \right)^{2}}}} & (17) \end{matrix}$

Starting from the corrected one-port system error parameters, a further secondary correction is implemented according to the invention by means of two-port and/or multiple-port measurements. In the following section, this is described by way of example for a two-port network analyzer. Generalization for further ports is made in an analogous manner by considering all VNA ports respectively in pairs.

The signal flow diagram of the 12-term error model of a vectorial two-port network analyzer illustrated in FIG. 5, which is expanded by comparison with the one-port measurement by six system parameters, is initially considered. Accordingly, in the forward direction, this involves the load-port match e₂₂, the transmission tracking (e₁₀e₃₂) and the crosstalk e₃₀. In the reverse direction (FIG. 6), corresponding parameters are used. It is assumed that the crosstalkers in forward and reverse direction are negligible, which, without restricting generality, is fulfilled for modern network analyzers. After performing a two-port calibration by means of a standard 12-term calibration method (for example, SOLT) or by means of an electronic calibration unit, the determination of the effective directivity and effective source-port match and the determination of the effective reflection synchronization is initially performed as described above. With knowledge of the effective one-port system parameters, values for the system parameters at every port, corrected according to equations (15)-(17), are calculated and, following this, overwritten into the corresponding memory registers of the network analyzer. By contrast, the two system parameters in transmission remain initially unchanged.

With the network analyzer having now been subjected to a partial secondary-error correction, a further measurement is performed, and in fact, that of an ideal through-connection of the two test ports. The result is a two-port scattering matrix

$\begin{matrix} {\left\lbrack S_{T} \right\rbrack = \left\lbrack {\begin{matrix} S_{T,11} \\ S_{T,\; 21} \end{matrix},\begin{matrix} S_{T,\; 12} \\ S_{T,\; 22} \end{matrix}} \right\rbrack} & (18) \end{matrix}$

from which, with the partially corrected system error set, the un-corrected scattering matrix or raw-value scattering matrix [S_(T,raw)] is calculated, which is defined as follows:

$\begin{matrix} {\left\lbrack S_{T,{raw}} \right\rbrack = \begin{bmatrix} \frac{b_{0}}{a_{0}} & \frac{b_{0}^{\prime}}{a_{3}^{\prime}} \\ \frac{b_{3}}{a_{0}} & \frac{b_{3}^{\prime}}{a_{3}^{\prime}} \end{bmatrix}} & \left( {18a} \right) \end{matrix}$

From the demand that the measurement of the through-connection in the case of a perfectly system-error-corrected network analyzer, should give the theoretical values, that is to say S_(T,11)=S_(T,22)=0 and S_(T,21)=S_(T,12)=1, the corrected system parameters for the load-port match and transmission tracking can be calculated according to

$\begin{matrix} {{\overset{\sim}{e}}_{22} = \frac{S_{T,{raw},11} - {\overset{\sim}{e}}_{00}}{{{\overset{\sim}{e}}_{10}{\overset{\sim}{e}}_{01}} + {{\overset{\sim}{e}}_{11}\left( {S_{T,{raw},11} - {\overset{\sim}{e}}_{00}} \right)}}} & (19) \\ {\left( {{\overset{\sim}{e}}_{10}{\overset{\sim}{e}}_{32}} \right) = {{S_{T,{raw},21}\left( {1 - {{\overset{\sim}{e}}_{11}{\overset{\sim}{e}}_{22}}} \right)}.}} & (20) \end{matrix}$

The determination of the corrected parameters in the reverse direction is performed in a corresponding manner. It should be noted that no explicit determination of the effective system parameters in the forward direction is performed, but a direct correction of the absolute system parameters “load-port match” and “transmission tracking”. By measuring all through-connections of a multi-port network analyzer, of which the one-port parameters have previously been subjected to a secondary-error correction, a complete correction can be performed by applying equations (19) and (20).

As an alternative, the further secondary error correction can be implemented on the basis of an 8-term error model (FIG. 7) using the UOSM method (also referred to as “unknown thru” method). This is a simplified error model, in which the load-port match is explicitly known. By way of difference from the 12-term SOLT calibration method, instead of the measurement of a through-connection, the measurement of a (possibly unknown) reciprocal, two-port network is implemented here, from which the seventh model parameter is determined with known one-port parameters (directivity, source-port match and reflection synchronization). On the basis of the network topology, the relationship between the measured and the actual scattering parameters is established in a transmission-matrix formulation via the cascading of three 2-port matrices. The following relationship between the measured transmission matrix [T_(m)] and the transmission matrix [T] of the device under test is then obtained:

[T_(m)]=[T₁][T][T₂]  (21)

with

$\begin{matrix} {\left\lbrack T_{1} \right\rbrack = {{\frac{1}{e_{10}}\begin{bmatrix} {- \left( {{e_{00}e_{11}} + {e_{10}e_{01}}} \right)} & e_{00} \\ {- e_{11}} & 1 \end{bmatrix}} = {\frac{1}{e_{10}}\lbrack A\rbrack}}} & (22) \\ {\left\lbrack T_{2} \right\rbrack = {{\frac{1}{e_{32}}\begin{bmatrix} {- \left( {{e_{22}e_{33}} + {e_{32}e_{23}}} \right)} & e_{22} \\ {- e_{33}} & 1 \end{bmatrix}} = {{\frac{1}{e_{32}}\lbrack B\rbrack}.}}} & (23) \end{matrix}$

The matrix elements in equations (22) and (23) refer to the one-port parameters at the two ports, while the product (e₁₀e₃₂) is obtained from the measurement of the reciprocal two-port network. According to the invention, once again, a first secondary error correction of the reflection parameters is also performed at both test ports in a similar manner, and the corrected parameters are updated in the VNA. The missing parameter of the transmission tracking is then calculated from

$\begin{matrix} {\frac{1}{e_{10}e_{32}} = {\pm {\sqrt{\frac{{\det \lbrack A\rbrack}{\det \lbrack B\rbrack}}{\det\left\lbrack T_{m} \right\rbrack}}.}}} & (24) \end{matrix}$

Since only system parameters already subjected to a secondary correction are contained on the right-hand side of equation (24) alongside the measured transmission matrix of the unknown through-connection, the determination of the missing parameter is also performed with very high accuracy.

The complete re-correction of all system parameters performed by means of the method according to the invention leads to a reduction of the measurement error in measurements in forward and or reverse direction between test ports by reducing the residual error terms in the transmission direction. Regarding the value of these residual error terms, the user has no precise information after a VNA calibration performed in a regular manner; he must rely on the information from the manufacture of conventional or electronic calibration kits, wherein relatively coarse estimates are generally specified in this case. A further advantage of the method is that, through its application, the test engineer obtains information regarding the quality of the previously performed, conventional system calibration and therefore regarding the quality of the calibration kits used. This is of great importance, because, especially at high-frequencies, the description of the elements of a calibration kit is based only on simulation data and possibly models the real physical properties of the calibration standard in an erroneous or inadequate manner. Accordingly, the present method can also be regarded as a verification method and is furthermore suitable as a method for characterizing calibration standards more accurately. The latter point is important in the context of manufacturers' descriptions of electronic calibration units, in which the switching states of the unit are measured by means of a high-precision calibrated VNA. If the latter were to be subjected to a secondary calibration by means of the present method, this would achieve a reduced measurement uncertainty attainable with an electronic calibration unit.

FIG. 8 once again shows a simplified method flow chart. Initially, the vectorial network analyzer is subjected to primary calibration (Step S1). For this purpose, conventional or electronic calibration kits can be used. After the implementation of the primary calibration in step S1, preferably at all one-port networks of the network analyzer, their residual system-error parameters are determined (S2). After the determination of the residual system-error parameters, the corrected system-error parameters are calculated in step S3 and the corresponding memory registers of the vectorial network analyzer are overwritten. As a result, the vectorial network analyzer is subjected to a partial secondary error correction. Accordingly, a first secondary correction of the system error parameters, which relate to the one-port networks of the VNA, has been performed. The subsequent, second secondary correction can be performed according to the simplified method flow chart illustrated on the left or on the right in FIG. 8.

In the case of the measurement of an ideal through-connection of the respective test ports of the network analyzer, a two-port scattering matrix [S_(T)] is determined in step S11. For their part, the two one-port networks have already been subjected to a secondary correction as described above. From the two-port scattering matrix (S_(T)) and the system errors already partially corrected on the basis of the first secondary correction, an un-corrected scattering matrix [S_(T,raw)] is calculated in step S12 from the determined two-port scattering matrix. Following this, a correction of the absolute system parameters “load-port match” and “transmission tracking” is then performed directly in step S13 from the un-corrected scattering matrix [S_(T,raw)].

As an alternative, after the calculation of the corrected system-error parameters S3, a reciprocal two-port network and its transmission matrix [T_(m)] can be determined (step S21). According to an eight-term error model (UOSM-method), the transmission tracking is then determined (step S22).

The detailed method steps and the mathematical relationships have already been explained in detail above, and a repetition of this description will not be provided at this point.

Reference is made to the fact that the above description is based upon the assumption that, for the one-port networks, corrected system-error parameters for the directivity, the source-port match and the reflection synchronization are available or determined. However, the method according to the invention can be used, even if only one of these corrected system-error parameters is available or determined, wherein reductions in the quality of the results must then, of course, be made. 

1. A method for error correction of a vectorial network analyzer with at least two test ports, with the following method steps: performing a primary system calibration using a calibration kit, performing a first secondary error correction on the relevant one-port networks of the vectorial network analyzer, performing of a second secondary error correction, wherein, after the first secondary error correction of the at least two one-port networks and the determination of at least one of the system parameters directivity, source-port match and the reflection synchronization on the one-port networks, a second secondary error correction is performed, wherein, for this purpose, the at least two one-port networks are through connected in an ideal manner and, taking into consideration the system-error parameter(s) of the first secondary error correction for at least one frequency point of the test-frequency range, a scattering parameter matrix is measured, from this, an un-corrected scattering parameter matrix is determined and, from this, a corrected load-port match and/or a corrected transmission tracking in forward and/or in reverse direction is determined.
 2. The method according to claim 1, wherein the measurement of the scattering parameter matrix is performed for every frequency point of a measurement range.
 3. A method for the error correction of a vectorial network analyzer with at least one two-port network, with the following method steps: performing of a primary system calibration using a calibration kit, performing of a first secondary error correction on at least two one-port networks of the vectorial network analyzer, performing of a second secondary error correction wherein, after the first secondary error correction of the at least two one-port networks and the determination of at least one of the system parameters directivity, source-port match and the reflection synchronization on the one-port networks, a second secondary error correction is performed, wherein, for this purpose, a measurement is performed on a reciprocal two-port network and, for the determination of a corrected transmission tracking, a transmission matrix is determined from the measurement on the reciprocal two-port network.
 4. The method according to claim 1, wherein the effective directivity and the source-port match is initially determined for each one-port network from a frequency-dependent measurement of the complex reflection coefficient of an air line short-circuited at its end and, following this, the effective reflection synchronization is determined by measuring an offset short-circuit or short-circuit for each frequency point of the test-frequency range as a complex value on each one-port network.
 5. The method according to claim 2, wherein the effective directivity and the source-port match is initially determined for each one-port network from a frequency-dependent measurement of the complex reflection coefficient of an air line short-circuited at its end and, following this, the effective reflection synchronization is determined by measuring an offset short-circuit or short-circuit for each frequency point of the test-frequency range as a complex value on each one-port network.
 6. The method according to claim 3, wherein the effective directivity and the source-port match is initially determined for each one-port network from a frequency-dependent measurement of the complex reflection coefficient of an air line short-circuited at its end and, following this, the effective reflection synchronization is determined by measuring an offset short-circuit or short-circuit for each frequency point of the test-frequency range as a complex value on each one-port network. 